Decorrelation tolerant coherent radar altimeter

ABSTRACT

A pulsed coherent radar altimeter is described which employs a narrow band receiver and utilizes a novel digital coherent pulse generator. A coherent pulse radar transmits a pulse comprised of the sum of at least two phase related RF signals closely spaced in frequency. The phase shift due to platform motion and return surface irregularity of the return signal is approximately the same for each carrier. The receiver produces a signal representative of the difference of the two carriers which is substantially free of decorrelation effects, and which can be processed in a narrow band receiver to produce range information.

This is a continuation-in-part of application Ser. No. 713118, filed 18 March 1985, now abandoned.

BACKGROUND OF THE INVENTION

This invention relates to a coherent radar, and more particularly to an improved coherent radar altimeter.

As those skilled in the art will understand, coherent radar systems are sensitive to the phase consistency of the received echoes. Radar echoes from an extended surface, such as the surface of the earth, are such that, for a radar carried by a moving platform, the relative phase of the radio frequency (RF) pulses in a sequential pulse train rapidly decorrelate. For this reason use of coherent radar for aircraft altimeters has been heretofore impractical.

At the same time, as will also be appreciated by those skilled in the art, coherent systems offer advantages of consequence in terms of low peak transmitter power (enabling the use of solid state RF transmitters at C-band) and relative immunity to spurious signals. In particular, these advantages would be significant in a precision altimeter based upon coherent pulse operation. For example, operation to 50,000 feet and higher necessitates coherent operation in order to utilize solid state transmitters of reasonable cost. But prior art proposals for coherent pulsed radar altimeters have not been satisfactory. As mentioned above, return signal phase decorrelation caused by a Doppler shift as a result of platform motion and phase shifts due to topological irregularities within the return surface area has made coherent pulse altimeters using narrow bandwidth receivers impractical.

SUMMARY OF THE INVENTION

The object of this invention is a pulsed coherent radar altimeter which employs a narrow band receiver.

Another object of the invention is the provision of a novel digital coherent pulse generator for use in a narrow band coherent radar altimeter.

Briefly, this invention contemplates the provision of a coherent pulse radar which transmits a pulse comprised of the sum of at least two phase related RF signals closely spaced in frequency. The phase shift due to platform motion and return surface irregularity of the return signal is approximately the same for each carrier. The receiver produces a signal representative of the difference of the two carriers, which is substantially free of decorrelation effects, and which can be processed in a narrow band receiver using otherwise prior art techniques to produce the desired range information.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of one embodiment of a coherent pulsed altimeter constructed in accordance with the teaching of this invention.

FIG. 2 is a block diagram of a pulse generator for use in the system of FIG. 1.

FIG. 3 is a schematic block diagram of a novel pulse generation using digital techniques to generate a coherent radar pulse having two RF carriers for use in the system.

FIG. 4a through 4h are a series of block diagrams useful in explaining the operation of FIG. 3.

DETAILED DESCRIPTION OF THE INVENTION

Referring now to FIG. 1, a moving platform 12, such as an aircraft moving from right to left, carries a pair of antennas 14 and 16, respectively a transmitting and a receiving antenna. A coherent RF pulse generator is connected to the transmitter antenna 14.

By way of explanation, the transmitter may comprise a pair of continuous wave oscillators 18 and 22 generating signals at two closely adjacent RF carrier frequencies f₁ and f₂. The oscillators 18 and 22 should be maintained in a fixed phase relationship to one another a indicated by dash line connection 20. This may be accomplished by any suitable manner known to those skilled in the art. FIG. 2 illustrates one technique for making source 18 and 22 provide signals S₁ and S₂, respectively, which are coherent with respect to each other.

In FIG. 2, an RF coupler 21 and a automatic phase control circuit 23 cross-couple two closely adjacent high frequency signal sources 18 and 22 to produce two phase related signals: ##EQU1## Signals S₁ and S₂ are coupled as inputs to an RF signal adder 25 whose output is coupled to a pulse forming switch 26. The output signal to antenna 14 is a series of RF pulses with an RF carrier signal equal to the sum of two coherent signals S₁ and S₂ having RF frequency f₁ and f₂ respectively.

In practice, the problem of providing two satisfactory coherent carriers at reasonable costs is difficult to solve using conventional techniques, such as shown in FIGS. 1 and 2. For this reason the novel generator of FIG. 3 is preferred and will be described in detail following the description of FIG. 1.

The resultant radar output signal, produced by the circuit of FIGS. 1 and 2 is a series of high frequency radar pulses sampled from a coherent signal representable by:

Equation A

    K=cos(w.sub.1 t+φ.sub.1)+cos (w.sub.2 t+φ.sub.2)

where:

K=radar signal

w₁ =2πf₁

w₂ =2πf₂

φ₁ & φ₂ =phase angles

The radar signal output K is equivalent to the simultaneous sum of two coherent high frequency signals.

The reflected signal from surface 20 a illustrated in FIG. 1, such as the surface of the earth, is received by an antenna 16 and coupled to the input of a conventional range gate 28.

A mixer 32 mixes the output of range gate 28 with a suitably chosen carrier f_(LO) and the resultant signal is coupled to a pair of filters 34 and 36 centered respectively at the translated frequencies chosen for f₁ and f₂, and having a bandwidth equal to the anticipated Doppler span (i.e., range of frequency shift anticipated as a result of platform motion).

Another mixer 38 mixes the output signals from the filters 34 and 36. This combined signal is coupled to a narrow band filter 42 whose frequency is centered on the difference frequency between carrier signal frequencies f₁ and f₂, and whose bandwidth is commensurate with the information base band bandwidth. The output of this narrow band filter 42 is coupled to a suitable radar information detection and processing system 44, known in the art, from which range information is extracted.

In the operation of the invention, the wavelength of radio waves at a radio altimeter frequency of 4.3 GHz is 0.228 feet or 2.7 inches. The amplitude correlation distance δ is dependent on the RF wavelength and has been estimated as:

Equation B ##EQU2## λ=RF wavelength v=ground speed

h=altitude above surface

τ=pulse width

C=velocity of light

By transmitting two coherent RF carrier signals of closely spaced frequency, simultaneously, for each pulse, the effective RF wavelength is artificially increased. Upon reflection from the surface and reception by the receiver in two IF-channels, the two signals are mixed together and the altitude tracker operates on the resulting difference frequency signal whose phase variation with distance is diminished by the factor:

Equation C ##EQU3##

For Example, at frequencies ##EQU4##

the phase rate factor is

13.9×10⁻⁶

FIG. 3 illustrates a novel, digitally constructed, two coherent carrier, phase coded, signal source in accordance with the present invention. Shown in FIG. 3 is a single RF oscillator 52 whose output is fed through a digitally driven phase inverting switch 54. The digitally driven switch 54 is driven by a pair of phase related pulse generators 56 and 58.

Generator 56 produces a continuous pulse train signal P₁ having pulse repetition frequency f_(p1), and generator 58 produces continuous pulse train signal P₂ having pulse repetition frequency f_(p2). Pulse signals P₁ and P₂ are graphically illustrated in FIG. 4c and 4d respectively. The phase related signals P₁ and P₂ may be constructed by a variety of techniques including digital counters and dividers driven by and synchronized with a common clock signal.

The output of digitally controlled switch 54 is a signal having a pair of phase related signal components at frequencies f_(p1) and f_(p2) with a controllable relative phase. Signals P₁ and P₂ are presented to a digital multiplier 65 for generating a product signal M which will be subsequently described. Signal M provides the control signal for digitally driven modulator switch 54.

Switch 54 is known as a bi-phase modulator. Switch 54 may be constructed by a variety of tehniques including appropriate application and control of a mixer as taught in RF SIGNAL PROCESSING COMPONENTS CATALOG of Watkins--Johnson Co., 1985/86, p. 66. In FIG. 3, switch 54 is illustrated using simply a pair of transmission gates 501 and 502. The output signal of RF oscillator 52, having frequency f_(c) is passed through transmission gate 501 upon a positive logic state signal M. The output of transmission gate 501 is presented to a summing circuit 503. Further, the output of RF oscillator 52 is presented to transmission gate 502 through a phase inverting circuit 505. Transmission gate 502 is controlled by signal M passed through an inverting circuit 507. The output of transmission gate 502 is presented as a second signal to summing circuit 503. The signal from oscillator 52 will reach output 62 either via switch 501 or 502, never both at the same time, as determined by signal M. The output of summing circuit 503 is indicated as signal 62. The output signal 62, as will be subsequently described, is substantially the same as the output of adder circuit 25 illustrated in FIGS. 1 and 2. To understand the operation of the circuit of FIG. 3, reference again is made to the circuit of FIG. 1 and its RF signal envelope as illustrated in FIG. 4a.

The resultant signal spectrum of the radar signal generator of FIG. 1 has frequency components at frequencies f₁ and f₂ with a frequency difference therebetween directly related to the frequency difference between the sources 18 and 22. The desired radar signal having frequency components at frequencies f₁ and f₂ is again represented by

Equation A, namely:

    K=cos(w.sub.1 t+φ.sub.1)+cos(w.sub.2 t+φ.sub.2)

This may be converted via trigonometric manipulation into:

Equation D ##EQU5##

FIG. 4a illustrates the envelope of radar signal K, in accordance with the present invention, where the radar signal is comprised of the sum of phase related carrier signals of differing frequency. FIG. 4a shows the envelope of radar signal K excluding any pulsing caused by switch 26. The exemplary waveform illustrated in FIG. 4a is one in which the ratio of the two frequencies of the two signals is six to one, and where the phase of the two signals is

    φ.sub.1 =φ.sub.2 =0.

Radar signal K, as will be readily appreciated from the form of Equation D, can be generated by the product modulation of two signals having frequencies (f₁ +f₂)/2 and (f₁ -f₂)/2, respectively.

FIG. 4b illustrates the conversion of the signal envelope FIG. 4a into a switched waveform. In FIG. 4b, the switch point phase of FIG. 4a is preserved but the amplitude is held constant. FIG. 4b therefore represents a phase modulation control signal M:

    M=signum (K).

The desired radar signal illustrated in FIG. 4a is synthesized in the present invention by phase modulating an RF carrier signal in accordance with the polarity of the phase modulation signal M or another signal indicative thereof. It will be appreciated by those skilled in the art that the waveform of FIG. 4b is the product of the waveforms shown in FIG. 4c and 4d which are representative of pulse train signals P₁ and P₂.

Referring again to FIG. 3, digital pulse generators 56 and 58 are presented to digital multiplier 65 to digitally construct the digital equivalent of modulation signal M. One example of digital multiplier 65 is illustrated in FIG. 3 as an exclusive-or circuit 67. The output of digital multiplier 67, signal M, is presented to bi-phase modulating switch 54.

In operation, switch circuit 54 of FIG. 3: (i) passes the signal of RF oscillator 52 to switch output 62 in response to one state of the output of digital multiplier 65, and (ii) passes 180° phase inverted signal of RF oscillator 52 to switch output 62 in response to a second state of the output of digital multiplier 65. The resulting signal output 62 is a signal substantially equivalent as that provided by the output of summing circuit 25 of FIG. 2. That is, the output radar signal of the circuit of FIG. 3 is a radar signal comprised of the sum of two phase related signals with different frequencies as shown in FIGS. 4b, c and d, but translated in frequency to be centered around the frequency f_(c) of oscillator 52 in FIG. 3.

Investigation of the spectrum produced by the circuit of FIG. 3 produces the line spectrum shown in FIG. 4e which has a desired frequency line pair a and a' plus a host of harmonics at least 9.4 dB down from the desired pair. The line spacing is

    f=f.sub.p1 -f.sub.p2

and the center frequency f_(c) is the frequency of RF signal oscillator 52.

The phase of the main signal components (a and a') is taken as {0, 0}or {180°, 180°} in synthesizing the radar signal resulting from the modulation signal of FIG. 4b. The specific choice of 0° or 180° is arbitrary but must be consistent once chosen. With two major carriers there are, then, four relative phase combinations. This may be diagrammatically shown as in FIG. 4f, where 0 means in-phase, and 1 means 180° out of phase.

With use of pulse generators 56 and 58, the phase relationship between the desired carrier signals having frequency having frequency difference f_(p2) -f_(p1) can be easily controlled. The switch drive functions shown in FIG. 4g can be easily generated from a reference pulse generator through standard logic elements, digital counters, flip-flops, and the like. Addition of a phase select switch allows sequential, individual, phase coding of a synthesized pair of signal carriers as determined by the pulse generator frequencies f_(p1) and f_(p2).

In applications where the host of sidebands shown in FIG. 4e is deleterious to the intended use, a sinusoidal taper may be applied as indicated in FIG. 4h.

Those skilled in the art will recognize that only the preferred embodiment of the present invention is disclosed herein and that the embodiment may be altered and modified without departing from the true spirit and scope of the invention as defined in the accompanying claims. 

What is claimed is:
 1. A coherent radar altimeter system comprising in combination:signal generating means for generating a series of coherent radar pulse signals having a high frequency carrier in which said high frequency carrier is comprised of two co-existing high frequency signal components of a first and a second frequency, respectively, closely spaced in frequency; means to transmit said series of coherent radar pulses; means for receiving return signal pulses, said return signal pulses being said transmitted pulses reflected from a target surface; means for processing said return signal pulses, including means for producing from said return signal pulses a signal which is representative of the difference between first and second return signal components of said returned signal pulses in which said first return signal component is of said first frequency and said second return signal component is of said second frequency.
 2. A coherent radar altimeter system as in claim 1 wherein said means for processing said return signal includes a pair of filters respectively centered at said first and second frequencies, a mixer, means to couple the output of each of said filters to said mixer, a narrow band filter, and means to couple the output of said mixer to said narrow band filter.
 3. A coherent radar altimeter system as in claim 2 wherein said narrow band filter has a prediction bandwidth equal to the information bandwidth.
 4. A coherent radar altimeter system as in claim 3 wherein said narrow band filter has a center frequency substantially equal to the frequency difference between said first and second frequencies.
 5. The coherent radar altimeter of claim 1 wherein said signal generating means includes:a high frequency oscillator providing a first signal; a controllable bi-phase modulating switch having input means for receiving said first signal, and a control input means for receiving a control signal, and an output means; a pair of phase related pulse generators for providing first and second pulse trains; and means for obtaining a product signal representative of the product of said first and second pulse trains, said product signal providing said control signal.
 6. A coherent radar altimeter system comprising in combination:a signal generator for generating an RF signal comprised of two phase related frequency components, said signal generator including,a high frequency oscillator for generating a first RF signal, a pair of phase related pulse generators for providing first and second pulse trains; first means responsive to said first and second pulse train signals for providing a product signal representative of the product of said first and second pulse train signals, and a controllable bi-phase modulating switch having input means connected to the output of said high frequency oscillator, and an output means, said switch including control means for receiving said product signal, said switch means operative to pass said first RF signal to said output means in response to a first condition of said first product signal, and passing said first RF signal to said output means with 180° phase inversion in response to a second condition of said product signal; means to transmit said series of coherent radar pulses; means for receiving return signals from said transmitted pulses from a target surface; and means for processing said return pulses, including means for producing from said return pulses a signal which is representative of the difference between first and second return signal components of said returned signal pulses in which said first return signal component is of said first frequency and said second return signal component is of said second frequency. 